Method and apparatus for canceling spread-spectrum noise

ABSTRACT

A spread-spectrum noise canceller (182) is provided. A received phase and a received amplitude for a first (216) and a second (236) component of a received spread-spectrum signal (200) is determined. The second component (236) is a replica of the first component (216) which was received at a different time. In addition, the spread-spectrum signal (200) includes a known signal. A portion of a spread-spectrum noise signal in the received signal (200) is canceled by generating an estimated signal (270) by spreading (260) the known signal at the second component received phase (204) with the known signal at the first component received phase (224) and adjusting a gain (268) of an integrated form of the spread known signal as a function of the received amplitudes of the first (216) and the second (236) components. Subsequently, the known signal is processed out of the received spread-spectrum signal (200) by subtracting (166) the estimated signal (270) from a demodulated form (216, 236) of the received spread-spectrum signal (200).

FIELD OF THE INVENTION

The present invention relates to communication systems which employspread-spectrum signals and, more particularly, to a method andapparatus for canceling spread-spectrum noise in a communicationchannel.

BACKGROUND OF THE INVENTION

In general, the purpose of a communication system is to transmitinformation-bearing signals from a source, located at one point, to auser destination, located at another point some distance away. Acommunication system generally consists of three basic components:transmitter, channel, and receiver. The transmitter has the function ofprocessing the message signal into a form suitable for transmission overthe channel. This processing of the message signal is referred to asmodulation. The function of the channel is to provide a physicalconnection between the transmitter output and the receiver input. Thefunction of the receiver is to process the received signal so as toproduce an estimate of the original message signal. This processing ofthe received signal is referred to as demodulation.

Two types of channels exist, namely, point-to point channels andbroadcast channels. Examples of point-to-point channels includeswirelines (e.g., local telephone transmission), microwave links, andoptical fibers. In contrast, broadcast channels provide a capabilitywhere many receiving stations may be reached simultaneously from asingle transmitter (e.g., local television and radio stations).

Analog and digital transmission methods are used to transmit a messagesignal over a communication channel. The use of digital methods offersseveral operational advantages over analog methods, including but notlimited to: increased immunity to channel noise and interference,flexible operation of the system, common format for the transmission ofdifferent kinds of message signals, and improved security ofcommunication through the use of encryption.

These advantages are attained at the cost of increased transmission(channel) bandwidth and increased system complexity. Through the use ofvery large-scale integration (VLSI) technology a cost-effective way ofbuilding the hardware has been developed.

One digital transmission method that may be used for the transmission ofmessage signals over a communication channel is pulse-code modulation(PCM). In PCM, the message signal is sampled, quantized, and thenencoded. The sampling operation permits representation of the messagesignal by a sequence of samples taken at uniformly spaced instants oftime. Quantization trims the amplitude of each sample to the nearestvalue selected from a finite set of representation levels. Thecombination of sampling and quantization permits the use of a code(e.g., binary code) for the transmission of a message signal. Otherforms of digital transmission use similar methods to transmit messagesignals over a communication channel.

When message signals are digitally transmitted over a band-limitedchannel, a form of interference known as intersymbol interference mayresult. The effect of intersymbol interference, if left uncontrolled, isto severely limit the rate at which digital data may be transmittedwithout error over the channel. The cure for controlling the effects ofintersymbol interference may be controlled by carefully shaping thetransmitted pulse representing a binary symbol 1 or 0.

Further, to transmit a message signal (either analog or digital) over abandpass communication channel, the message signal must be manipulatedinto a form suitable for efficient transmission over the channel.Modification of the message signal is achieved by means of a processtermed modulation. This process involves varying some parameter of acarrier wave in accordance with the message signal in such a way thatthe message information is preserved and that the spectrum of themodulated wave contained in the assigned channel bandwidth.Correspondingly, the receiver is required to re-create the originalmessage signal from a degraded version of the transmitted signal afterpropagation through the channel. The re-creation is accomplished byusing a process known as demodulation, which is the inverse of themodulation process used in the transmitter.

In addition to providing efficient transmission, there are other reasonsfor performing modulation. In particular, the use of modulation permitsmultiplexing, that is, the simultaneous transmission of signals fromseveral message sources over a common channel. Also, modulation may beused to convert the message signal into a form less susceptible to noiseand interference.

Typically, in propagating through a channel, the transmitted signal isdistorted because of nonlinearities and imperfections in the frequencyresponse of the channel. Other sources of degradation are noise andinterference added to the received signal during the course oftransmission through the channel. Noise and distortion constitute twobasic limitations in the design of communication systems.

There are various sources of noise, internal as well as external to thesystem. Although noise is random in nature, it may be described in termsof its statistical properties such as the average power or the spectraldistribution of the average power.

In any communication system, there are two primary communicationresources to be employed, namely, average transmitted power and channelbandwidth. The average transmitted power is the average power of thetransmitted signal. The channel bandwidth defines the range offrequencies that the channel uses for the transmission of signals withsatisfactory fidelity. A general system design objective is to use thesetwo resources as efficiently as possible. In most channels, one resourcemay be considered more important than the other. Hence, we may alsoclassify communication channels as power-limited or band-limited. Forexample, the telephone circuit is a typical band-limited channel,whereas a deep-space communication link or a satellite channel istypically power-limited.

The transmitted power is important because, for a receiver of prescribednoise figure, it determines the allowable separation between thetransmitter and receiver. In other words, for a receiver of prescribednoise figure and a prescribed distance between it and the transmitter,the available transmitter power determines the signal-to-noise ratio atthe receiver input. This, subsequently, determines the noise performanceof the receiver. Unless this performance exceeds a certain design level,the transmission of message signals over the channel is not consideredto be satisfactory.

Additionally, channel bandwidth is important; because, for a prescribedband of frequencies characterizing a message signal, the channelbandwidth determines the number of such message signals that can bemultiplexed over the channel. In other words, for a prescribed number ofindependent message signals that have to share a common channel, thechannel bandwidth determines the band of frequencies that may beallotted to the transmission of each message signal without discernibledistortion.

For spread-spectrum communication systems, these areas of concern havebeen optimized in one particular manner. In spread-spectrum systems, amodulation technique is utilized in which a transmitted signal is spreadover a wide frequency band. The frequency band is wider than the minimumbandwidth required to transmit the information being sent. A voicesignal, for example, can be sent with amplitude modulation (AM) in abandwidth only twice that of the information itself. Other forms ofmodulation, such as low deviation frequency modulation (FM) or singlesideband AM, also permit information to be transmitted in a bandwidthcomparable to the bandwidth of the information itself. A spread-spectrumsystem, on the other hand, often takes a baseband signal (e.g., a voicechannel) with a bandwidth of only a few kilohertz, and distributes itover a band that may be many megahertz wide. This is accomplished bymodulating with the information to be sent and with a wideband encodingsignal. Through the use of spread-spectrum modulation, a message signalmay be transmitted in a channel in which the noise power is higher thanthe signal power. The modulation and demodulation of the message signalprovides a signal-to-noise gain which enables the recovery of themessage signal from a noisy channel. The greater the signal-to-noiseratio for a given system equates to: (1) the smaller the bandwidthrequired to transmit a message signal with a low rate of error or (2)the lower the average transmitted power required to transmit a messagesignal with a low rate of error over a given bandwidth.

Three general types of spread-spectrum communication techniques exist,including:

The modulation of a carrier by a digital code sequence bit rate is muchhigher than the information signal bandwidth. Such systems are referredto as "direct sequence" modulated systems.

Carrier frequency shifting in discrete increments in a pattern dictatedby a code sequence. These systems are called "frequency hoppers". Thetransmitter jumps from frequency to frequency within some predeterminedset; the order of frequency usage is determined by a code sequence.Similarly "time hopping" and "time-frequency hopping" have times oftransmission which are regulated by a code sequence.

Pulse-FM or "chirp" modulation in which a carrier is swept over a wideband during a given pulse interval.

Information (i.e., the message signal) can be embedded in the spectrumsignal by several methods. One method is to add the information to thespreading code before it is used for spreading modulation. Thistechnique can be used in direct sequence and frequency hopping systems.It will be noted that the information being sent must be in a digitalform prior to adding it to the spreading code, because the combinationof the spreading code, typically a binary code, involves modulo-2addition. Alternatively, the information or message signal may be usedto modulate a carrier before spreading it.

Thus, a spread-spectrum system must have two properties: (1) thetransmitted bandwidth should be much greater than the bandwidth or rateof the information being sent, and (2) some function other than theinformation being sent is employed to determine the resulting modulatedchannel bandwidth.

The essence of the spread-spectrum communication involves the art ofexpanding the bandwidth of a signal, transmitting the expanded signaland recovering the desired signal by remapping the receivedspread-spectrum into the original information bandwidth. Furthermore, inthe process of carrying out this series of bandwidth trades, the purposeof spread-spectrum techniques is to allow the system to deliverinformation with low error rates in a noisy signal environment.

The present invention enhances the ability of spread-spectrum systemsand, in particular, code division multiple access (CDMA) cellularradio-telephone systems to recover spread-spectrum signals from a noisyradio communication channel. In CDMA cellular radio-telephone systems,the "users" are on the same frequency and separated only by unique usercodes. The noise interference level in the communication channel isdirectly related to the interference level created by the users plusadditive Gaussian noise and not solely by additive Gaussian noise likein other communication systems. Thus, the number of users that cansimultaneously use the same frequency band in a given cellular regionwith a low relative of additive Gaussian noise is limited primarily bythe code noise of all active "users". The present invention reduces theeffects of undesired user code noise and thus significantly increasesthe number of users which can simultaneously be serviced by a givencellular region.

SUMMARY OF THE INVENTION

A spread-spectrum noise canceller is provided. A received phase and areceived amplitude for a first and a second component of a receivedspread-spectrum signal are determined. The second component is a replicaof the first component which was received at a different time. Inaddition, the spread-spectrum signal includes a known signal. A portionof a spread-spectrum noise signal in the received signal is canceled bygenerating an estimated signal by spreading the known signal at thesecond component received phase with the known signal at the firstcomponent received phase and adjusting a gain of an integrated form ofthe spread known signal as a function of the received amplitudes of thefirst and the second components. Subsequently, the known signal isprocessed out of the received spread-spectrum signal by subtracting theestimated signal from a demodulated form of the received spread-spectrumsignal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing a prior art spread-spectrum communicationsystem.

FIG. 2 is a diagram showing a preferred embodiment internal structure ofa receiver having a spread-spectrum noise canceller for use in the priorart spread-spectrum communication system shown in FIG. 1.

FIG. 3 is a flowchart summarizing the operation of the preferredembodiment noise canceller shown in FIG. 2.

DETAILED DESCRIPTION

Referring now to FIG. 1, a prior art spread-spectrum communicationsystem, as substantially described in U.S. Pat. No. 5,103,459 forGilhousen et al. filed Jun. 25,1990, and "On the System Design Aspectsof code Division Multiple Access (CDMA) Applied to Digital Cellular andPersonal Communication Networks," Allen Salmasi and Klein S. Gilhousen,presented at the 41st IEEE Vehicular Technology Conference on May 19-22,1991 in St. Louis, Mo., pages 57-62, is shown.

In the prior art spread-spectrum communication system, traffic channeldata bits 100 are input to an encoder 102 at a particular bit rate(e.g., 9.6 kilobits/second). The traffic channel data bits can includeeither voice converted to data by a vocoder, pure data, or a combinationof the two types of data. Encoder 102 convolutionally encodes the inputdata bits 100 into data symbols at a fixed encoding rate. For example,encoder 102 encodes received data bits 100 at a fixed encoding rate ofone data bit to two data symbols such that the encoder 102 outputs datasymbols 104 at a 19.2 kilosymbols/second rate. The encoder 102accommodates the input of data bits 100 at variable rates by encodingrepetition. That is when the data bit rate is slower than the particularbit rate at which the encoder 102 is designed to operate, then theencoder 102 repeats the input data bits 100 such that the input databits 100 are provided to the encoding elements within the encoder 102 atthe equivalent of the input data bit rate at which the encoding elementsare designed to operate. Thus, the encoder 102 outputs data symbols 104at the same fixed rate regardless of the rate at which data bits 100 areinput to the encoder 102.

The data symbols 104 are then input into an interleaver 106. Interleaver106 interleaves the input data symbols 104. This interleaving of relateddata symbols 104 causes bursts of errors in a communication channel 138to be spread out in time and thus to be handled by a decoder 178 as ifthey were independent random errors. Since, communication channel 138memory decreases with time separation, the idea behind interleaving isto separate (i.e., make independent) the related data symbols 104 of anencoded data bit 100 in time. The intervening space in a transmissionblock is filled with other data symbols 104 related to other encodedbits 100. Separating the data symbols 104 sufficiently in timeeffectively transforms a communication channel 138 with memory into amemoryless one, and thereby enables the use of the random-errorcorrecting codes (e.g., convolutional codes and block codes).Subsequently, a maximum likelihood convolutional decoder 178 can make adecision based on a sequence of data samples 176 of a received signal inwhich each data sample 176 is assumed to be independent from the otherdata samples 176. Such an assumption of independence of data samples 176or memorylessness of the communication channel 138 can improve theperformance of a maximum likelihood decoder 178 over a decoder whichdoes not make such assumptions. The interleaved data symbols 108 areoutput by the interleaver 106 at the same data symbol rate that theywere input (e.g., 19.2 kilosymbols/second) to one input of anExclusive-OR/multiplier 112.

A long pseudo-noise (PN) generator 110 is operatively coupled to theother input of the Exclusive-OR/multiplier 112 to enhance the securityof the communication in the communication channel by scrambling the datasymbols 108. The long PN generator 110 uses a long PN sequence togenerate a user specific sequence of symbols or unique user spreadingcode at a fixed rate equal to the data symbol rate of the data symbols108 which are input to the other input of the Exclusive-OR gate 112(e.g., 19.2 kilosymbols/second). The scrambled data symbols 114 areoutput from the Exclusive-OR/multiplier 112 at a fixed rate equal to therate that the data symbols 108 are input to the Exclusive-OR gate 112(e.g., 19.2 kilosymbols/second) to one input of anExclusive-OR/multiplier 118.

A code division channel selection generator 116 provides a particularpredetermined length Walsh code to the other input of theExclusive-OR/multiplier 118. The code division channel selectiongenerator 116 can provide one of 64 orthogonal codes corresponding to 64Walsh codes from a 64 by 64 Hadamard matrix wherein a Walsh code is asingle row or column of the matrix. The Exclusive-OR/multiplier 118 usesthe particular Walsh code input by the code division channel generator116 to spread the input scrambled data symbols 114 into Walsh codespread data symbols 120. It will be appreciated by those skilled in theart that "spreading" is a term used to describe the operation ofincreasing the number of symbols which represent input data symbols. Forexample, the combiner 118 may receive a sequence of scrambled datasymbols 114 at a rate of 19.2 kilosymbols/second. Each scrambled datasymbol 114 is combined with a Walsh spreading code 116 such that eachscrambled data symbol 114 is represented by or spread into a single 64bit length Walsh spreading code 120. As a result, the Walsh code spreaddata symbols 120 are output from the Exclusive-OR/multiplier 118 at afixed chip rate (e.g., 1.2288 Megachips/second). The term "chip" is usedin the art interchangeable with the term "bits" when describing segmentsof a spread digital signal.

The Walsh code spread data symbols 120 are provided to an input of twoExclusive-OR/multipliers 122 and 128, respectively. A pair of short PNsequences (i.e., short when compared to the long PN sequence used by thelong PN generator 110) are generated by I-channel PN generator 124 andQ-channel PN generator 130. These PN generators 124 and 130 may generatethe same or different short PN sequences. The Exclusive-OR/multipliers122 and 128 further spread the input Walsh code spread data 120 with theshort PN sequences generated by the PN I-channel generator 124 and PNQ-channel generator 130, respectively. The resulting I-channel codespread sequence 126 and Q-channel code spread sequence 132 are used toquadrature-phase shift key (QPSK) modulate a quadrature pair ofsinusoids 134 by driving the power level controls of a pair ofsinusoids. The sinusoids' output signals are summed, bandpass filtered,translated to a radio frequency (RF), amplified, filtered and radiatedby an antenna 136 to complete transmission of the traffic channel databits 100 in a communication channel 138.

Antenna 140 receives a spread-spectrum signal such that the receivedsignal can be processed in a substantially complementary set ofoperations as compared to the set of operations performed on the trafficchannel data bits 100 prior to their transmission over communicationchannel 138 by antenna 136. The received spread-spectrum signal istranslated to a baseband frequency, filtered, and QPSK demodulated 142into a demodulated spread-spectrum signal 144, 146. Subsequently, thedemodulated spread-spectrum signal 144, 146 is quadrature despread. Apair of short PN sequences are generated by I-channel PN generator 148and Q-channel PN generator 154. These PN generators 148 and 154 mustgenerate the same short PN sequences as the PN generators 124 and 130,respectively. The Exclusive-OR/multipliers 150 and 152 despread theinput demodulated spread-spectrum signals 144 and 146, respectively. Theresulting I-channel code despread sequence 156 and Q-channel codedespread sequence 158 are combined into quadrature despread data samples160.

A code division channel selection generator 164 provides a particularpredetermined length Walsh code to an input of theExclusive-OR/multiplier 162. The code division channel selectiongenerator 164, like generator 116, can provide one of 64 orthogonalcodes corresponding to 64 Walsh codes from a 64 by 64 Hadamard matrixwherein a Walsh code is a single row or column of the matrix, but toproperly despread a particular code transmission the same Walsh code asthe transmitter generator 116 generated must be generated. TheExclusive-OR/multiplier 162 uses the particular Walsh code input by thecode division channel generator 164 to despread the input quadraturedespread data samples 160 into Walsh code despread data samples 166. Itwill be appreciated by those skilled in the art that "despreading" is aterm used to describe the operation of decreasing the number of sampleswhich represent input. For example, the combiner 162 may receive asequence of despread data samples 160 at a rate of 1.2288Megasamples/second. A group of 64 despread data samples 160 is combinedwith a selected Walsh despreading code 164 such that the group of 64despread data samples 160 is represented by or despread into a singleWalsh despread data sample 166. As a result, the Walsh code despreaddata samples 166 are output from the Exclusive-OR/multiplier 162 at afixed rate (e.g., 19.2 kilosamples/second).

A long PN generator 170 is operatively coupled to the input of theExclusive-OR/multiplier 168 to descramble the despread data samples 166.The long PN generator 170 uses a long PN sequence to generate a userspecific sequence of samples or unique user spreading code at a fixedrate equal to the data samples rate of the despread data samples 166which are input to the other input of the Exclusive-OR gate 168 (e.g.,19.2 kilosamples/second). This operation uses the same long PN sequenceas generated by long PN generator 110 and is the logical complement ofthe scrambling operation performed by the Exclusive-OR gate 112. Thedescrambled data samples 172 are output from the Exclusive-OR/multiplier168 at a fixed rate equal to the rate that the despread data samples 166are input to the Exclusive-OR gate 168 (e.g., 19.2 kilosamples/second).

The descrambled data samples 172 are then input into a deinterleaver174. Deinterleaver 174 deinterleaves the input descrambled data samples172 in a manner which is the logical complement of the interleaver 106.The deinterleaved data samples 176 are output by the deinterleaver 174at the same data sample rate that they were input (e.g., 19.2kilosamples/second). Subsequently, a maximum likelihood convolutionaldecoder 178 makes decisions based on the input sequence of deinterleaveddata samples 176. The maximum likelihood decoder 178 preferablygenerates estimated data bits 180 by utilizing maximum likelihooddecoding techniques which are substantially similar to the Viterbidecoding algorithm.

Referring now to FIG. 2, a diagram is shown of a preferred embodimentinternal structure of a portion 182 of a receiver having aspread-spectrum noise canceller for use in the prior art spread-spectrumcommunication system shown in FIG. 1.

The receiver portion 182 as described hereinafter preferably isimplemented in a mobile communication unit of a cellular radiocommunication system also having a plurality of base stations or centralcommunication sites. It will be appreciated by those skilled in the artthat the particular receiver portion structure 182 having a noisecanceller described herein could readily be adapted for use in thecentral communication sites or in any other communication system havingsimilar knowledge of the multipath characteristics of the signalreceived on a communication channel.

It will be appreciated by those skilled in the art that spreading codesother than Walsh spreading codes 116, 164 can be used to separate datasignals from one another in a CDMA communication system. For instance,PN spreading codes can be used to separate a plurality of data signals.A particular data signal can be separated from the other data signals byusing a particular PN spreading code which is offset by a particularphase to spread the particular data signal. For example, in a CDMAspread-spectrum communication system, a particular PN spreading code canbe used to generate a plurality of channels by using a different offsetphase for the PN spreading code for each channel of the communicationsystem.

Furthermore, the modulation scheme of the signals is assumed to bequadrature phase shift keying (QPSK). However, it will be appreciated bythose skilled in the art that other modulation techniques can be usedwithout departing from the teachings of the present invention. Finally,in the preferred embodiment, the communication channel 138 for thecellular communication system is in the 900 MHz region of theelectromagnetic spectrum. However, other regions of the electromagneticspectrum may be used without departing from the teachings of the presentinvention.

The portion 182 of the receiver shown implements "Rake" receivingtechniques to reduce the effect of multipath fading in the communicationchannel. It will be appreciated by those skilled in the art that "Rake"receiving techniques are well known in the art of radio communication.For example, "A Communication Technique for Multipath Channels," R.Price and P. E. Green, Jr., Proceedings of the IRE, March 1958, pages555-570 describes the basic operation of a "Rake" receiver. Briefly, a"Rake" receiver performs a continuous, detailed measurement of themultipath characteristic of a received signal. This knowledge is thenexploited to combat the selective fading by detecting the echo signalsindividually, using a correlation method, and algebraically combiningthose echo signals into a single detected signal. The intersymbolinterference is attenuated by varying the time delay or phase betweenthe various detected echo signals prior to their algebraic combination.

Similar to the prior art communication system shown in FIG. 1, theantenna 140, shown in FIG. 2 receives a spread-spectrum signal such thatthe received signal can be processed in a substantially complementaryset of operations as compared to the set of operations performed on thetraffic channel data bits 100 prior to their transmission overcommunication channel 138 by antenna 136. The received spread-spectrumsignal is a composite signal including several signals in differentspread-spectrum channels. At least one of these spread-spectrum signalsis a known pilot data signal. Each of spread-spectrum signals in thecomposite received spread-spectrum signal may be received by receiver182 from one or more base stations and along one or more communicationpaths. As a result, each of the signals in a particular spread-spectrumchannel may have several components which vary in amplitude and/or phasefrom the other signals in the channel. In the preferred embodiment,similar pilot data signals are transmitted from each base station in thecommunication system. However, when a mobile communication unit isattempting to retrieve (i.e., demodulate and decode) a particular signalfrom a spread-spectrum channel, these pilot data signals contribute tothe non-deterministic noise in the communication channel 138. Theseundesired signals can be canceled when the receiver has obtainedparticular information concerning the communication channel and thereceived composite spread-spectrum signal.

The spread-spectrum signal received on antenna 140 is translated to abaseband frequency, filtered, and QPSK demodulated 142 into ademodulated spread-spectrum signal 200. During this demodulation process142, a received phase and a received amplitude for each component of thereceived spread-spectrum signal is determined. The phase represents themoment in time that a particular component is received relative to theother components. The amplitude represents the relative received signalstrength or received accuracy of the component relative to the othercomponents. During the following discussion, the receivedspread-spectrum signal is assumed to have a signal in one particularspread-spectrum channel and further that the signal has threecomponents. These signal components have followed differentcommunication channel paths on their way to receiver 182. For thisexample, the first component was transmitted by a base station in aprimary serving cell and was received at a phase φ₁ and an amplitude A₁.Similarly, the second component was transmitted by the base station inthe primary serving cell, but traveled along a different communicationpath than the first component, and was received at a phase φ₂ and anamplitude A₂ . Finally, the third component was transmitted by a basestation in a secondary serving cell (e.g., during a soft hand-offsituation) and was received at a phase φ₃ and an amplitude A₃.

In the preferred embodiment "Rake" receiver 182, demodulatedspread-spectrum signal 200 is input to individual receiver portionswhich manipulate each of the three signal components. The first signalcomponent is quadrature despread by inputting the demodulatedspread-spectrum signal 200 into Exclusive-OR combiner 202. A pair ofshort PN sequences are generated by I-channel PN generator 148 andQ-channel PN generator 154 (shown in FIG. 1). The pair of short PNsequences is input 204 to the Exclusive-OR combiner 202 at the firstcomponent phase φ₁. Exclusive-OR combiner 202 despreads the inputdemodulated spread-spectrum signal 200. In addition, Exclusive-ORcombiner 202 combines the resulting I-channel code despread sequence andQ-channel code despread sequence into quadrature despread data samples206. It will be appreciated by those skilled in the art that although asingle Exclusive-OR combiner 202 is described above, like in the priorart receiver, shown in FIG. 1, two Exclusive-OR/multipliers (e.g.,multipliers 150 and 152) could be used.

The quadrature despread data samples 206 for the first signal componentare input to Exclusive-OR/multiplier 208. A code division channelselection generator 164 (shown in FIG. 1) provides a particularpredetermined length Walsh code (W_(i)) at the first signal componentphase φ₁ 210 to the other input of the Exclusive-OR/multiplier 208. TheExclusive-OR/multiplier 208 uses the particular Walsh code (W_(i)) 210input by the code division channel generator 164 to despread the inputquadrature despread data samples 206 into Walsh code despread datasamples 212.

These Walsh code despread data samples 212 are then input to integrator214 which integrates the data samples 212 over a predetermined timeperiod (T) and adjusts the gain of the input data samples 212 signal.The predetermined time period (T) preferably corresponds to a desiredoutput rate of data samples from the "Rake" receiver 182 (e.g., 19.2kilosamples/second output rate which corresponds to T=1/19,200 of asecond). The gain of the input data samples 212 signal is adjusted by again factor g₁ which is a function of the amplitude of the first signalcomponent A₁ (g₁ =f(A₁)). This gain factor g₁ is also determined suchthat it enables maximum ratio combining of the three signal components.In addition, the input data sample 212 gain is divided/adjusted by thepredetermined time period (T) so that the output signal 216 gain betterreflects the gain associated with each input data sample 212. The outputof integrator 214 is a Walsh code despread data sample signal 216 forthe first signal component. This first signal component Walsh codedespread data sample signal 216 may optionally be switched into an input218 of a signal processor 220. It will be appreciated by those skilledin the art that the integrator 214 function may be implemented with adata sample summing circuit and multiplier.

The second signal component can be derived from the demodulatedspread-spectrum signal 200 in a manner similar to that previouslydescribed for the first signal component. The second signal component isquadrature despread by inputting the demodulated spread-spectrum signal200 into Exclusive-OR combiner 222. A pair of short PN sequences aregenerated by I-channel PN generator 148 and Q-channel PN generator 154(shown in FIG. 1). The pair of short PN sequences is input 224 to theExclusive-OR combiner 222 at the second component phase φ₂. Exclusive-ORcombiner 222 despreads the input demodulated spread-spectrum signal 200and combines the resulting I-channel code despread sequence andQ-channel code despread sequence into quadrature despread data samples226.

The quadrature despread data samples 226 for the second signal componentare input to Exclusive-OR/multiplier 228. A code division channelselection generator 164 (shown in FIG. 1) provides a particularpredetermined length Walsh code (W_(i)) at the second signal componentphase φ₂ to the other input of the Exclusive-OR/multiplier 228. TheExclusive-OR/multiplier 228 uses the particular Walsh code (W_(i)) atthe second signal component phase φ₂ 230 input by the code divisionchannel generator 164 to despread the input quadrature despread datasamples 226 into Walsh code despread data samples 232.

These Walsh code despread data samples 232 are then input to integrator234 which integrates the data samples 232 over a predetermined timeperiod (T) and adjusts the gain of the input data samples 232 signal.The gain of the input data samples 232 signal is adjusted by a gainfactor g₂ which is a function of the amplitude of the second signalcomponent A₂ (g₂ =f(A₂)). This gain factor g₂ is also determined suchthat it enables maximum ratio combining of the three signal components.In addition, the input data sample 232 gain is divided/adjusted by thepredetermined time period (T) so that the output signal 236 gain betterreflects the gain associated with each input data sample 232. The outputof integrator 234 is a Walsh despread data sample signal 236 for thesecond signal component. This second signal component Walsh codedespread data sample signal 236 may optionally be switched into an input238 of the signal processor 220.

The third signal component can be derived from the demodulatedspread-spectrum signal 200 in a manner similar to that previouslydescribed for the first and second signal components. The third signalcomponent is quadrature despread by inputting the demodulatedspread-spectrum signal 200 into Exclusive-OR combiner 242. A pair ofshort PN sequences are generated by I-channel PN generator 148 andQ-channel PN generator 154 (shown in FIG. 1). The pair of short PNsequences is input 244 to the Exclusive-OR combiner 242 at the thirdcomponent phase φ₃. Exclusive-OR combiner 242 despreads the inputdemodulated spread-spectrum signal 200 and combines the resultingI-channel code despread sequence and Q-channel code despread sequenceinto quadrature despread data samples 246.

The quadrature despread data samples 246 for the third signal componentare input to Exclusive-OR/multiplier 248. A code division channelselection generator 164 (shown in FIG. 1) provides a particularpredetermined length Walsh code (W_(j)) at the third signal componentphase φ₃ to the other input of the Exclusive-OR/multiplier 248. TheExclusive-OR/multiplier 248 uses the particular Walsh code (W_(j)) atthe third signal component phase φ₃ 250 input by the code divisionchannel generator 164 to despread the input quadrature despread datasamples 246 into Walsh code despread data samples 252.

These Walsh code despread data samples 252 are then input to integrator254 which integrates the data samples 252 over a predetermined timeperiod (T) and adjusts the gain of the input data samples 252 signal.The gain of the input data samples 252 signal is adjusted by a gainfactor g₃ which is a function of the amplitude of the third signalcomponent A₃ (g₃ =f(A₃)). This gain factor g₃ is also determined suchthat it enables maximum ratio combining of the three signal components.In addition, the input data sample 252 gain is divided/adjusted by thepredetermined time period (T) so that the output signal 256 gain betterreflects the gain associated with each input data sample 252. The outputof integrator 254 is a Walsh despread data sample signal 256 for thethird signal component. This third signal component Walsh code despreaddata sample signal 256 may optionally be switched into an input 258 ofthe signal processor 220.

The demodulated spread-spectrum signal 200 further includesnon-deterministic noise consisting of two components. The two componentsto the non-deterministic noise are:

All of the CDMA spread-spectrum signals which are not being demodulatedby the receiver. These consist of a large number of low-levelinterfering users using the same communication channel as the receiverwhich are in nearby cells of the communication system.

Receiver front end noise. By design, additive noise preferably is belowthe demodulated spread-spectrum signal 200 when the communicationchannel is operating at full capacity.

A portion of this first spread-spectrum noise component can be canceledfrom the demodulated spread-spectrum signal 200 provided sufficientinformation is known to the receiver. This information includes severalpieces of data already known to a typical "Rake" receiver like thepreferred embodiment receiver portion 182 described above. This knowndata includes: the amplitude (i.e. A₁, A₂, and A₃) and phase (i.e., φ₁,φ₂, and φ₃) of each signal component, the short PN spread code sequences148 and 154 used by the communication system, and the Walsh code (W_(i))for the particular channel being received. With this known data, thereceiver portion 182 may be configured to cancel the noise related toother signal components such as a pilot channel carrier signal which maybe interfering with the desired signal components.

Typically each Walsh code channel does not contribute any noise to theother Walsh code channels because orthogonality is maintained. However,this is not true when there is significant delay spread (≧one chipdelay) and/or when the receiving unit is in a communication channelhand-off state between two or more transmitters. A possible situation inwhich these other channels may contribute noise or cause interference inthe desired communication channel is when either a delayed replica ofthe transmitted carrier or transmitted carriers of originating in othercells is received in the desired communication channel by the receiverportion 182 and the receiver portion 182 does not distinguish betweenthe desired signal and the interfering signals. As more of theseinterfering signals contribute to the demodulated spread-spectrum signal200 received by the receiver, the signal to noise ratio may deteriorateto near or below a preferred threshold.

In the preferred embodiment communication system, the delayed pilotsignal replicas of the primary serving cell and pilot signal energy fromother nearby cells cause approximately 1 dB of the total noise in thedesired communication channel. Through the following cancellationprocess, most of that 1 dB of noise can be canceled which results in agreater signal to noise ratio for the desired signal. Some of theadvantages of this cancellation technique include: removing or reducingundesired pilot channel signal interference from the received signal andallowing an increase in the number of users on a particular CDMAcommunication channel due to the increased capability of the receiversto handle interference in the communication channel.

A first estimated interference signal can be derived from the knowndata. The previously generated pair of short PN sequences having asecond component phase φ₂ are input 224 to an Exclusive-OR combiner 260.Similarly, previously generated pair of short PN sequences having afirst component phase φ₁ are input 204 to an Exclusive-OR combiner 260.Exclusive-OR combiner 260 spreads the second component phase φ₂sequences 224 with the first component phase φ₁ sequences 204 andcombines the resulting I-channel code spread sequence and Q-channel codespread sequence into quadrature spread data samples 262.

The quadrature spread data samples 262 for the first estimatedinterference signal are input to Exclusive-OR/multiplier 264. Thepreviously generated particular predetermined length Walsh code (W_(i))at the first signal component phase φ₁ 210 is provided to the otherinput of the Exclusive-OR/multiplier 264. The Exclusive-OR/multiplier264 uses the particular Walsh code (W_(i)) at the first signal componentphase φ₁ 210 to spread the input quadrature spread data samples 262 intoWalsh code spread data samples 266.

These Walsh code spread data samples 266 are then input to integrator268 which integrates the data samples 266 over a predetermined timeperiod (T) and adjusts the gain of the input data samples 266 signal.The gain of the input data samples 266 signal is adjusted by a negativeof a product of gain factor g₁ and g₂ (-g₁ ·g₂) which as previouslynoted are functions of the amplitudes of the first and second signalcomponents A₁ and A₂, respectively. This gain factor -g₁ ·g₂ is alsodetermined such that it enables a subtraction from the maximum ratiocombination of the three signal components (i.e., a negative factor). Inaddition, the input data sample 266 gain is adjusted by thepredetermined time period (T) so that the output signal 270 gain betterreflects the gain associated with each input data sample 266. The outputof integrator 268 is a first estimated Walsh despread data sampledinterference signal 270. This first estimated interference signal 270may optionally be switched into or input to 272 a signal processor 220.

A second estimated interference signal can also be derived from theknown data. The previously generated pair of short PN sequences having athird component phase φ₃ are input 244 to an Exclusive-OR combiner 280.Similarly, previously generated pair of short PN sequences having afirst component phase φ₁ are input 204 to an Exclusive-OR combiner 280.Exclusive-OR combiner 280 spreads the third component phase φ₃ sequences244 with the first component phase φ₁ sequences 204 and combines theresulting I-channel code spread sequence and Q-channel code spreadsequence into quadrature spread data samples 282.

The quadrature spread data samples 282 for the second estimatedinterference signal are input to Exclusive-OR/multiplier 284. Thepreviously generated particular predetermined length Walsh code (W_(j))at the first signal component phase φ₁ 210 is provided to the otherinput of the Exclusive-OR/multiplier 284. The Exclusive-OR/multiplier264 uses the particular Walsh code (W_(i)) at the first signal componentphase φ₁ 210 to spread the input quadrature spread data samples 282 intoWalsh code spread data samples 286.

These Walsh code spread data samples 286 are then input to integrator288 which integrates the data samples 286 over a predetermined timeperiod (T) and adjusts the gain of the input data samples 286 signal.The gain of the input data samples 286 signal is adjusted by a negativeof a product of gain factor g₁ and g₃ (-g₁ ·g₃) which as previouslynoted are functions of the amplitudes of the first and third signalcomponents A₁ and A₃, respectively. This gain factor -g₁ ·g₃ is alsodetermined such that it enables a subtraction from the maximum ratiocombination of the three signal components (i.e., a negative factor). Inaddition, the input data sample 286 gain is adjusted by thepredetermined time period (T) so that the output signal 290 gain betterreflects the gain associated with each input data sample 286. The outputof integrator 288 is a second estimated Walsh despread data sampledinterference signal 290. This second estimated interference signal 290may optionally be switched into or input to 292 a signal processor 220.

The generation of the first and second estimated interference signal wasmade by way of example only. It will be appreciated by those skilled inthe art that this estimated interference signal process may be continuedfor any other interfering signal for which sufficient information isknown.

Finally, signal processor 220 preferably maximum ratio combines severalsignal components (e.g., signal components 216, 236, 256, 270, and/or290) into a single Walsh code despread data sample 166 signal. Thissingle Walsh code despread data sample 166 signal is output from thesignal processor 220 at a fixed rate (e.g., 19.2 kilosamples/second).Subsequently, the Walsh code despread data sample 166 signal ispreferably further processed in a manner similar to the prior artreceiver, shown in FIG. 1, to generate estimated data bits 180.

It will be appreciated by those skilled in the art that it may not bedesirable to cancel all of the interfering signals from the desiredsignal. Thus, the signal strengths of the interfering signals may becompared to the desired signal. Further, only the undesired interferingsignals having a signal strength greater than the desired signal shouldbe removed from the composite demodulated spread-spectrum signal 200. Ifthe weaker undesired interfering signals are removed, then the desireddata signal may be partially corrupted. In addition, it will beappreciated by those skilled in the art that a spread-spectrum signal(e.g., the desired signal) typically can be detected and retrieved froma composite signal when it's signal strength is greater than the signalstrengths of interfering signals. Thus, the removal of interferingsignals from composite signal which have a signal strength less than thedesired signal is unnecessary and may unduly increase the detection andretrieval time of the desired signal.

For example, in the case of the desired signal having the three signalcomponent Walsh code despread data sample signals 216, 236, and 256, aninterferer is removed from the combined single Walsh code despread datasample 166 signal, if it has a stronger signal strength than the desiredsignal components. For instance, the first estimated signal 270 may havea signal strength greater than the third signal component Walsh codedespread data sample signal 256 and as such should be removed from thecombined single Walsh code despread data sample 166 signal. Thus, thefirst estimated signal 270 is switched input 272 of the signal processor220. In contrast, the second estimated signal 290 may have a signalstrength less than the third signal component Walsh code despread datasample signal 256 and as such should not be removed from the combinedsingle Walsh code despread data sample 166 signal. Thus, the secondestimated signal 290 is not switched input 292 of the signal processor220. It will be appreciated by those skilled in the art that anothersignal quality or communication system metric may be used to determinewhich interfering signals should be canceled from the compositespread-spectrum signal without departing from the teachings of thepresent invention. For example, the cancellation of particularinterfering signals may be determined by a comparison of a predeterminedthreshold to a function of the adjusted gains (g₁, g₂, and g₃) of theintegrators 214, 234, and 254, respectively.

The operation of the preferred embodiment noise canceller is summarizedas the flowchart shown in FIG. 3. A spread-spectrum signal having afirst and a second component is received 300 from over a communicationchannel. The first component being received at a different time from thesecond component. In addition, the received spread-spectrum signalincludes a known signal (e.g., a cellular communication system pilotchannel signal). A received phase (φ₁, φ₂) and a received amplitude (A₁,A₂) for the first and the second components of a receivedspread-spectrum signal is determined 302, respectively. Subsequently,the received spread-spectrum signal first and second components aredemodulated and maximum ratio combined 304. In addition, an estimatedsignal is generated 306 by spreading the known signal at the secondcomponent received phase (φ₂) with the known signal at the firstcomponent received phase (φ₁), spreading the known signal at the secondcomponent received phase (φ₂) with a channel selecting spreading code atthe first component received phase (φ₁), integrating the spread knownsignal over a predetermined time (T), and adjusting a gain of theintegrated spread known signal as a function of the received amplitudesof the first (A₁) and the second components (A₂). Subsequently, aportion of a spread-spectrum noise signal is canceled 310 by processingthe known signal out of the received spread-spectrum signal throughsubtracting the estimated signal from the demodulated receivedspread-spectrum signal only if 308 the adjusted gain of the integratedform of the spread known signal (g₁ g₂) is greater than a predeterminedthreshold. Subsequently, the spread-spectrum signal receiving processmay be completed by descrambling 312 the processed demodulated form ofthe received spread-spectrum signal by utilizing a known spreading code.In addition, the descrambled received spread-spectrum signal isdeinterleaved 314 within a predetermined size block. Finally, at leastone estimated data bit is generated 316 by utilizing maximum likelihooddecoding techniques which are substantially similar to the Viterbidecoding algorithm to derive the at least one estimated data bit fromthe deinterleaved received spread-spectrum signal.

Although the invention has been described and illustrated with a certaindegree of particularity, it is understood that the present disclosure ofembodiments has been made by way of example only and that numerouschanges in the arrangement and combination of parts as well as steps maybe resorted to by those skilled in the art without departing from thespirit and scope of the invention as claimed. For example, it will beappreciated by those skilled in the art that the above described noisecancellation techniques can be performed in the intermediate or basebandfrequencies without departing from the spirit and scope of the presentinvention as claimed. In addition, the modulator, antennas anddemodulator portions of the preferred embodiment communication system asdescribed were directed to spread-spectrum signals transmitted over aradio communication channel. However, as will be understood by thoseskilled in the art, the communication channel could alternatively be anelectronic data bus, wireline, optical fiber link, or any other type ofcommunication channel.

What is claimed is:
 1. An apparatus comprising a spread-spectrum noisecanceller, the spread-spectrum noise canceller comprising:(a)determining means for determining a received phase and a receivedamplitude for a first and a second component of a receivedspread-spectrum signal, the first component being received at adifferent time from the second component, the spread-spectrum signalincluding a known signal; and (b) noise canceling means, operativelycoupled to the determining means, for canceling a portion of aspread-spectrum noise signal in the received spread-spectrum signalby:(i) generating an estimated signal by spreading the known signal atthe second component received phase with the known signal at the firstcomponent received phase and adjusting a gain of an integrated form ofthe spread known signal as a function of the received amplitudes of thefirst and the second components; and (ii) processing the known signalout of the received spread-spectrum signal by subtracting the estimatedsignal from a demodulated form of the received spread-spectrum signal.2. The apparatus of claim 1 wherein the spread-spectrum noise cancellercanceling means processes the known signal out of the receivedspread-spectrum signal only if the adjusted gain of the integrated formof the spread known signal is greater than a predetermined threshold. 3.The apparatus of claim 1 wherein the spread-spectrum noise cancellercanceling means generates the estimated signal by further spreading theknown signal at the second component received phase with a channelselecting spreading code at the first component received phase.
 4. Theapparatus of claim 1 further comprising:(a) descrambling means,operatively coupled to the spread-spectrum noise canceller, fordescrambling the processed demodulated form of the receivedspread-spectrum signal by utilizing a known spreading code; (b)deinterleaving means, operatively coupled to the descrambling means, fordeinterleaving the descrambled received spread-spectrum signal within apredetermined size block; and (c) decoding means, operatively coupled tothe deinterleaving means, for generating at least one estimated data bitby utilizing maximum likelihood decoding techniques to derive the atleast one estimated data bit from the deinterleaved receivedspread-spectrum signal.
 5. The apparatus of claim 4 wherein the decodingmeans comprises means for generating at least one estimated data bit byutilizing maximum likelihood decoding techniques which are substantiallysimilar to the Viterbi decoding algorithm.
 6. An apparatus comprising aspread-spectrum noise canceller, the spread-spectrum noisecomprising:(a) receiving means for receiving a spread-spectrum signalhaving a first and a second component from over a communication channel,the first component being received at a different time from the secondcomponent, the spread-spectrum signal including a known signal; (b)determining means, operatively coupled to the receiving means, fordetermining a received phase and a received amplitude for the first andthe second components of a received spread-spectrum signal; (c)demodulating means, operatively coupled to the receiving means, fordemodulating the received spread-spectrum signal; and (d) noisecanceling means, operatively coupled to the determining means and thedemodulating means, for canceling a portion of a spread-spectrum noisesignal in the received spread-spectrum signal by:(i) generating anestimated signal by spreading the known signal at the second componentreceived phase with the known signal at the first component receivedphase, integrating the spread known signal over a predetermined time,and adjusting a gain of the integrated spread known signal as a functionof the received amplitudes of the first and the second components; and(ii) processing the known signal out of the received spread-spectrumsignal by subtracting the estimated signal from the demodulated receivedspread-spectrum signal.
 7. The apparatus of claim 6 wherein thespread-spectrum noise canceller canceling means processes the knownsignal out of the received spread-spectrum signal only if the adjustedgain of the integrated form of the spread known signal is greater than apredetermined threshold.
 8. The apparatus of claim 6 wherein thespread-spectrum noise canceller canceling means generates an estimatedsignal by further spreading the known signal at the second componentreceived phase with a channel selecting spreading code at the firstcomponent received phase.
 9. The apparatus of claim 6 furthercomprising:(a) descrambling means, operatively coupled to thespread-spectrum noise canceller, for descrambling the processeddemodulated form of the received spread-spectrum signal by utilizing aknown spreading code; (b) deinterleaving means, operatively coupled tothe descrambling means, for deinterleaving the descrambled receivedspread-spectrum signal within a predetermined size block; and (c)decoding means, operatively coupled to the deinterleaving means, forgenerating at least one estimated data bit by utilizing maximumlikelihood decoding techniques to derive the at least one estimated databit from the deinterleaved received spread-spectrum signal.
 10. Theapparatus of claim 6 wherein the communication channel is selected fromthe group consisting essentially of an electronic data bus, radiocommunication link, wireline and optical fiber link.
 11. Aspread-spectrum signal processing method, comprising:(a) determining areceived phase and a received amplitude for a first and second componentof a received spread-spectrum signal, the first component being at adifferent time from the second component, the spread-spectrum signalincluding a known signal; and (b) canceling a portion of aspread-spectrum noise signal in the received spread-spectrum signalby:(i) generating an estimated signal by spreading the known signal atthe second component received phase with the known signal at the firstcomponent received phase and adjusting a gain of an integrated form ofthe spread known signal as a function of the received amplitudes of thefirst and the second components; and (ii) processing the known signalout of the received spread-spectrum signal by subtracting the estimatedsignal from a demodulated form of the received spread-spectrum signal.12. The method of claim 11 wherein the step of canceling comprisesprocessing the known signal out of the received spread-spectrum signalonly if the adjusted of the integrated form of the spread known signalis greater than a predetermined threshold.
 13. The method of claim 11wherein the step of canceling comprises generating the estimated signalby further spreading the known signal at the second component receivedphase with a channel selecting spreading code at the first componentreceived phase.
 14. The method of claim 11 further comprising:(a)descrambling the processed demodulated form of the receivedspread-spectrum signal by utilizing a known spreading code; (b)deinterleaving the descrambled received spread-spectrum signal within apredetermined size block; and (c) generating at least one estimated databit by utilizing maximum likelihood decoding techniques to derive the atleast one estimated data bit from the deinterleaved receivedspread-spectrum.
 15. The method of claim 14 wherein the step ofgenerating at least one estimated data bit comprises utilizing maximumlikelihood decoding techniques which are substantially similar to theViterbi decoding algorithm.
 16. A spread-spectrum signal processingmethod, comprising:(a) receiving a spread-spectrum signal having a firstand a second component from over a communication channel, the firstcomponent being received at a different time from the second component,the spread-spectrum signal including a known signal; (b) determining areceived phase and a received amplitude for the first and the secondcomponents of a received spread-spectrum signal; (c) demodulating thereceived spread-spectrum signal; and (d) canceling a portion of aspread-spectrum noise signal in the received spread-spectrum signalby:(i) generating an estimated signal by spreading the known signal atthe second component received phase with the known signal at the firstcomponent received phase, integrating the spread known signal over apredetermined time, and adjusting a gain of the integrated spread knownsignal as a function of the received amplitudes of the first and thesecond components; and (ii) processing the known signal out of thereceived spread-spectrum signal by subtracting the estimated signal fromthe demodulated received spread-spectrum signal.
 17. The method of claim16 wherein the step of canceling comprises processing the known signalout of the received spread-spectrum signal only if the adjusted gain ofthe integrated form of the spread known signal is greater than apredetermined threshold.
 18. The method of claim 16 wherein the step ofcanceling comprises generating the estimated signal by further spreadingthe known signal at the second component received phase with a channelselecting spreading code at the first component received phase.
 19. Themethod of claim 16 further comprising:(a) descrambling the processeddemodulated form of the received spread-spectrum signal by utilizing aknown spreading code; (b) deinterleaving the descrambled receivedspread-spectrum signal within a predetermined size block; and (c)generating at least one estimated data bit by utilizing maximumlikelihood decoding techniques to derive the at least one estimated databit from the deinterleaved received spread-spectrum.
 20. The method ofclaim 19 wherein the communication channel is selected from the groupconsisting essentially of an electronic data bus, radio communicationlink, wireline and optical fiber link.